UNLESS OTHERWISE NOTED, this document contains PRODUCTION DATA.
Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality.
The TLV7011 and TLV7021 are micro-power comparators with reasonable response time. The comparators have a rail-to-rail input stage that can monitor signals beyond the positive supply rail with integrated hysteresis. When higher levels of hysteresis are required, positive feedback can be externally added. The push-pull output stage of the TLV7011 is optimal for reduced power budget applications and features no shoot-through current. When level shifting or wire-ORing of the comparator outputs is needed, the TLV7021 with its open-drain output stage is well suited to meet the system needs. In either case, the wide operating voltage range, low quiescent current, and micro-package of the TLV7011 and TLV7021 make these comparators excellent candidates for battery-operated and portable, handheld designs.
The inverting comparator with hysteresis requires a three-resistor network that is referenced to the comparator supply voltage (VCC), as shown in Figure 38. When VIN at the inverting input is less than VA, the output voltage is high (for simplicity, assume VO switches as high as VCC). The three network resistors can be represented as R1 || R3 in series with R2. Equation 1 defines the high-to-low trip voltage (VA1).
When VIN is greater than VA, the output voltage is low, very close to ground. In this case, the three network resistors can be presented as R2 || R3 in series with R1. Use Equation 2 to define the low to high trip voltage (VA2).
Equation 3 defines the total hysteresis provided by the network.
A noninverting comparator with hysteresis requires a two-resistor network, as shown in Figure 39, and a voltage reference (VREF) at the inverting input. When VIN is low, the output is also low. For the output to switch from low to high, VIN must rise to VIN1. Use Equation 4 to calculate VIN1.
When VIN is high, the output is also high. For the comparator to switch back to a low state, VIN must drop to VIN2 such that VA is equal to VREF. Use Equation 5 to calculate VIN2.
The hysteresis of this circuit is the difference between VIN1 and VIN2, as shown in Equation 6.
Window comparators are commonly used to detect undervoltage and overvoltage conditions. Figure 40 shows a simple window comparator circuit.
For this design, follow these design requirements:
Configure the circuit as shown in Figure 40. Connect VCC to a 3.3-V power supply and VEE to ground. Make R1, R2 and R3 each 10-MΩ resistors. These three resistors are used to create the positive and negative thresholds for the window comparator (VTH+ and VTH–). With each resistor being equal, VTH+ is 2.2 V and VTH- is 1.1 V. Large resistor values such as 10-MΩ are used to minimize power consumption. The sensor output voltage is applied to the inverting and noninverting inputs of the two TLV7021's. The TLV7021 is used for its open-drain output configuration. Using the TLV7021 allows the two comparator outputs to be Wire-Ored together. The respective comparator outputs will be low when the sensor is less than 1.1 V or greater than 2.2 V. VOUT will be high when the sensor is in the range of 1.1 V to 2.2 V.
A single TLV7011 device can be used to build a complete IR receiver analog front end (AFE). The nanoamp quiescent current and low input bias current make it possible to be powered with a coin cell battery, which could last for years.
For this design, follow these design requirements:
The IR receiver AFE design is highly streamlined and optimized. R1 converts the IR light energy induced current into voltage and applies to the inverting input of the comparator. Because a reverse biased IR LED is used as the IR receiver, a higher I/V transimpedance gain is required to boost the amplitude of reduced current. A 10M resistor is used as R1 to support a 1-V, 100-nA transimpedance gain. This is made possible with the picoamps Input bias current IB (5pA typical). The RC network of R2 and C1 establishes a reference voltage Vref which tracks the mean amplitude of the IR signal. The RC constant of R2 and C1 (about 4.7 ms) is chosen for Vref to track the received IR current fluctuation but not the actual data bit stream. The noninverting input is connected to Vref and the output over the R3 and R4 resistor network which provides additional hysteresis for improved guard against spurious toggles.
To reduce the current drain from the coin cell battery, data transmission must be short and infrequent.
Square-wave oscillator can be used as low cost timing reference or system supervisory clock source.
The square-wave period is determined by the RC time constant of the capacitor and resistor. The maximum frequency is limited by propagation delay of the device and the capacitance load at the output. The low input bias current allows a lower capacitor value and larger resistor value combination for a given oscillator frequency, which may help to reduce BOM cost and board space.
The oscillation frequency is determined by the resistor and capacitor values. The following calculation provides details of the steps.
First consider the output of Figure Figure 44 is high which indicates the inverted input VC is lower than the noninverting input (VA). This causes the C1 to be charged through R4, and the voltage VC increases until it is equal to the noninverting input. The value of VA at the point is calculated by Equation 7.
if R1 = R2= R3, then VA1 = 2 VCC/ 3
At this time the comparator output trips pulling down the output to the negative rail. The value of VAat this point is calculated by Equation 8.
if R1 = R2 = R3, then VA2 = VCC/3
The C1 now discharges though the R4, and the voltage VCC decreases until it reaches VA2. At this point, the output switches back to the starting state. The oscillation period equals to the time duration from for C1 from 2VCC/3 to VCC / 3 then back to 2VCC/3, which is given by R4C1 × ln 2 fro each trip. Therefore, the total time duration is calculated as 2 R4C1 × ln 2. The oscillation frequency can be obtained by Equation 9:
Figure 46 shows the simulated results of tan oscillator using the following component values: