Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality.
The OPA835 and OPA2835 devices can be used as noninverting amplifiers with signal input to the noninverting input, VIN+. A basic block diagram of the circuit is shown in Figure 53.
If VIN = VREF + VSIG, the amplifier output may be calculated according to Equation 1.
The signal gain of the circuit is set by , and VREF provides a reference around which the input and output signals swing. Output signals are in-phase with the input signals.
The OPA835 and OPA2835 devices are designed for the nominal value of RF to be 2 kΩ in gains other than +1. This gives excellent distortion performance, maximum bandwidth, best flatness, and best pulse response. RF = 2 kΩ must be used as a default unless other design goals require changing to other values. All test circuits used to collect data for this data sheet had RF = 2 kΩ for all gains other than +1. A gain of +1 is a special case where RF is shorted and RG is left open.
The OPA835 and OPA2835 devices can be used as inverting amplifiers with signal input to the inverting input, VIN–, through the gain-setting resistor RG. A basic block diagram of the circuit is shown in Figure 54.
If VIN = VREF + VSIG, the output of the amplifier may be calculated according to Equation 2.
The signal gain of the circuit and VREF provides a reference point around which the input and output signals swing. Output signals are 180˚ out-of-phase with the input signals. The nominal value of RF must be 2 kΩ for inverting gains.
Figure 64 is an instrumentation amplifier that combines the high input impedance of the differential-to-differential amplifier circuit and the common-mode rejection of the differential-to-single-ended amplifier circuit. This circuit is often used in applications where high input impedance is required (such as taps from a differential line) or in cases where the signal source is a high impedance.
If VIN+ = VCM + VSIG+ and VIN– = VCM + VSIG–, the output of the amplifier may be calculated according to Equation 3.
The signal gain of the circuit is . VCM is rejected, and VREF provides a level shift around which the output signal swings. The single-ended output signal is in-phase with the differential input signal.
Integrated solutions are available, but the OPA835 device provides a much lower-power, high-frequency solution. For best CMRR performance, resistors must be matched. A good rule of thumb is CMRR ≈ the resistor tolerance; so 0.1% tolerance will provide approximately 60-dB CMRR.
The noninverting circuit shown in Figure 53 has a minimum gain of 1. To implement attenuation, a resistor divider can be placed in series with the positive input, and the amplifier set for a gain of 1 by shorting VOUT to VIN– and removing RG. Because the op amp input is high impedance, the resistor divider sets the attenuation.
The inverting circuit of Figure 54 is used as an attenuator by making RG larger than RF. The attenuation is the resistor ratio. For example, a 10:1 attenuator can be implemented with RF = 2 kΩ and RG = 20 kΩ.
Figure 65 shows an amplifier circuit that converts single-ended signals to differential signals and provides gain and level shifting. This circuit can convert signals to differential in applications such as driving Cat5 cabling or driving differential-input SAR and ΔΣ ADCs.
By setting VIN = VREF + VSIG, then the output of the amplifier may be calculated according to Equation 4.
The differential-signal gain of the circuit is 2 × G, and VREF provides a reference around which the output signal swings. The differential output signal is in-phase with the single-ended input signal.
Line termination on the output can be accomplished with resistors RO. The differential impedance seen from the line will be 2 × RO. For example, if 100-Ω Cat5 cable is used with double termination, the amplifier is typically set for a differential gain of 2 V/V (6 dB) with RF = 0 Ω (short) RG = ∞Ω (open), 2R = 2 kΩ, R1 = 0 Ω, R = 1 kΩ to balance the input bias currents, and RO = 49.9 Ω for output line termination. This configuration is shown in Figure 66.
For driving a differential-input ADC the situation is similar, but the output resistors, RO, are selected with a capacitor across the ADC input for optimum filtering and settling-time performance.
Figure 67 shows a differential amplifier that converts differential signals to single-ended and provides gain (or attenuation) and level shifting. This circuit can be used in applications like a line receiver for converting a differential signal from a Cat5 cable to a single-ended signal.
If VIN+ = VCM + VSIG+ and VIN– = VCM + VSIG–, then the output of the amplifier may be calculated according to Equation 5.
The signal gain of the circuit is , VCM is rejected, and VREF provides a level shift around which the output signal swings. The single-ended output signal is in-phase with the differential input signal.
Line termination can be accomplished by adding a shunt resistor across the VIN+ and VIN- inputs. The differential impedance is the shunt resistance in parallel with the input impedance of the amplifier circuit, which is usually much higher. For low gain and low line impedance, the resistor value to add is approximately the impedance of the line. For example, if a 100-Ω Cat5 cable is used with a gain of 1 amplifier and RF = RG = 2 kΩ, adding a 100-Ω shunt across the input will give a differential impedance of 99 Ω, which is adequate for most applications.
For best CMRR performance, resistors must be matched. Assuming CMRR ≈ the resistor tolerance, a 0.1% tolerance will provide about 60-dB CMRR.
Figure 68 shows a differential amplifier that is used to amplify differential signals. This circuit has high input impedance and is used in differential line driver applications where the signal source is a high-impedance driver (for example, a differential DAC) that must drive a line.
If the user sets VIN± = VCM + VSIG±, then the output of the amplifier may be calculated according to Equation 6.
The signal gain of the circuit is , and VCM passes with unity gain. The amplifier combines two noninverting amplifiers into one differential amplifier that shares the RG resistor, which makes RG effectively ½ its value when calculating the gain. The output signals are in-phase with the input signals.
The OPA835 RUN package option includes integrated gain-setting resistors for the smallest possible footprint on a printed circuit board (≈ 2.00 mm x 2.00 mm). By adding circuit traces on the PCB, gains of +1, –1, –1.33, +2, +2.33, -3, +4, –4, +5, –5.33, +6.33, –7, +8 and inverting attenuations of –0.1429, –0.1875, –0.25, –0.33, –0.75 can be achieved.
Figure 69 shows a simplified view of how the OPA835IRUN integrated gain-setting network is implemented. Table 3 lists the required pin connections for various noninverting and inverting gains (reference Figure 53 and Figure 54). Table 4 lists the required pin connections for various attenuations using the inverting-amplifier architecture (reference Figure 54). Due to ESD protection devices being used on all pins, the absolute maximum and minimum input voltage range, VS– – 0.7 V to VS+ + 0.7 V, applies to the gain-setting resistors, and so attenuation of large input voltages will require external resistors to implement.
The gain-setting resistors are laser trimmed to 1% tolerance with nominal values of 2.4 kΩ, 1.8 kΩ, and 600 Ω. The gain-setting resistors have excellent temperature coefficient, and gain tracking is superior to using external gain-setting resistors. The 800-Ω resistor and 1.25-pF capacitor in parallel with the 2.4-kΩ gain-setting resistor provide compensation for best stability and pulse response.
|SHORT PINS||SHORT PINS||SHORT PINS||SHORT PINS|
|1 V/V (0 dB)||—||1 to 9||—|
|2 V/V (6.02 dB)||–1 V/V (0 dB)||1 to 9||2 to 8||6 to GND||—|
|2.33 V/V (7.36 dB)||–1.33 V/V (2.5 dB)||1 to 9||2 to 8||7 to GND||—|
|4 V/V (12.04 dB)||–3 V/V (9.54 dB)||1 to 8||2 to 7||6 to GND||—|
|5 V/V (13.98 dB)||–4 V/V (12.04 dB)||1 to 9||2 to 7 or 8||7 to 8||6 to GND|
|6.33 V/V (16.03 dB)||–5.33 V/V (14.54 dB)||1 to 9||2 to 6 or 8||6 to 8||7 to GND|
|8 V/V (18.06 dB)||–7 V/V (16.90 dB)||1 to 9||2 to 7||6 to GND||—|
|SHORT PINS||SHORT PINS||SHORT PINS||SHORT PINS|
|–0.75 V/V (–2.5 dB)||1 to 7||2 to 8||9 to GND||—|
|–0.333 V/V (–9.54 dB)||1 to 6||2 to 7||8 to GND||—|
|–0.25 V/V (–12.04 dB)||1 to 6||2 to 7 or 8||7 to 8||9 to GND|
|–0.1875 V/V (–14.54 dB)||1 to 7||2 to 6 or 8||6 to 8||9 to GND|
|–0.1429 V/V (–16.90 dB)||1 to 6||2 to 7||9 to GND||—|
For pulsed applications where the signal is at ground and pulses to a positive or negative voltage, the circuit bias-voltage considerations differ from those in an application with a signal that swings symmetrically about a reference point.Figure 70 shows a circuit where the signal is at ground (0 V) and pulses to a positive value.
If the input signal pulses negative from ground, an inverting amplifier is more appropriate, as shown in Figure 71. A key consideration in noninverting and inverting cases is that the input and output voltages are kept within the limits of the amplifier. Because the VICR of the OPA835 device includes the negative supply rail, the OPA835 op amp is well-suited for this application.
The OPA835 device provides excellent performance when driving high-performance delta-sigma (ΔΣ) and successive-approximation-register (SAR) ADCs in low-power audio and industrial applications.
To show achievable performance, the OPA835 device is tested as the drive amplifier for the ADS8326 device. The ADS8326 device is a 16-bit, micro power, SAR ADC with pseudodifferential inputs and sample rates up to
250 kSPS. The device offers excellent noise and distortion performance in a small 8-pin SOIC or VSSOP (MSOP) package. Low power and small size make the ADS8326 and OPA835 devices an ideal solution for portable and battery-operated systems, remote data-acquisition modules, simultaneous multichannel systems, and isolated data acquisition.
|TONE (Hz)||SIGNAL (dBFS)||SNR (dBc)||THD (dBc)||SINAD (dBc)||SFDR (dBc)|
The OPA835 and OPA2835 devices provide excellent audio performance with low quiescent power. To show performance in the audio band, an audio analyzer from Audio Precision (2700 series) tests THD+N and FFT at 1 VRMS output voltage.
Figure 74 shows the test circuit used for the audio-frequency performance application.
Design a low distortion, single-ended input to single-ended output audio amplifier using the OPA835 device. The 2700 series audio analyzer from Audio Precision is the signal source and the measurement system.
|OPA835 Unity Gain Config.||1 KHz Tone Frequency||> 110 dBc SFDR||300 Ω and
The OPA835 device is tested in this application in a unity-gain buffer configuration. A buffer configuration is selected as the configuration maximizes the loop gain of the amplifier configuration. At higher closed-loop gains, the loop gain of the circuit reduces, which results in degraded harmonic distortion. The relationship between distortion and closed loop gain at a fixed input frequency can be seen in Figure 36 in Typical Characteristics: VS = 5 V. The test was performed under varying output-load conditions using a resistive load of 300 Ω and 100 KΩ. Figure 34 shows the distortion performance of the amplifier versus the output resistive load. Output loading, output swing, and closed-loop gain play a key role in determining the distortion performance of the amplifier.
The 100-pF capacitor to ground on the input helped to decouple noise pickup in the lab and improved noise performance.
The Audio Precision was configured as a single-ended output in this application circuit. In applications where a differential output is available, the OPA835 device can be configured as a differential to single-ended amplifier as shown in Figure 67. Power supply bypassing is critical to reject noise from the power supplies. A 2.2-μF power-supply decoupling capacitor must be placed within two inches of the device and can be shared with other op amps on the same board. A 0.1-μF decoupling capacitor must be placed as close to the power supply pins as possible, preferably within 0.1 inch. For split supply, a capacitor is required for both supplies. A 0.1-µF capacitor placed directly between the supplies is also beneficial for improving system noise performance. If the output load is heavy, from 16 Ω to 32 Ω, amplifier performance could begin to degrade. To drive such heavy loads, both channels of the OPA2835 device can be paralleled with the outputs isolated with 1-Ω resistors to reduce the loading effects.
A 10-Ω series resistor can be inserted between the capacitor and the noninverting pin to isolate the capacitance.
Figure 75 shows the THD+N performance with 100-kΩ and 300-Ω loads, and with no weighting and A-weighting. With no weighting, the THD+N performance is dominated by the noise for both loads. A-weighting provides filtering that improves the noise so a larger difference can be seen between the loads due to more distortion with RL = 300 Ω.
Figure 76 and Figure 77 show FFT output with a 1-kHz tone and 100-kΩ and 300-Ω loads. To show relative performance of the device versus the test set, one channel has the OPA835 device in-line between generator output and analyzer input, and the other channel is in “Gen Mon” loopback mode, which internally connects the signal generator to the analyzer input. With 100-kΩ load (see Figure 76), the curves are indistinguishable from each other except for noise, which means the OPA835 device cannot be directly measured. With a 300-Ω load as shown in Figure 77, the main difference between the curves is the OPA835 device due to the higher even-order harmonics. The test-set performance masks the odd-order harmonics.
The OPA835 and OPA2835 devices are good choices for active filters. Figure 79 and Figure 78 show MFB and Sallen-Key circuits designed using the WEBENCH® Filter Designer to implement second-order low-pass Butterworth filter circuits. Figure 80 shows the frequency response.
Other MFB and Sallen-Key filter circuits display similar performance. The main difference is the MFB is an inverting amplifier in the pass band and the Sallen-Key is noninverting. The primary advantage for each is the Sallen-Key in unity gain has no resistor gain error term, and thus no sensitivity to gain error, while the MFB has better attenuation properties beyond the bandwidth of the op amp.