SLVS497F SEPTEMBER   2003  – June 2016 TPS65140 , TPS65141 , TPS65145


  1. Features
  2. Applications
  3. Description
  4. Revision History
  5. Pin Configuration and Functions
  6. Specifications
    1. 6.1 Absolute Maximum Ratings
    2. 6.2 ESD Ratings
    3. 6.3 Recommended Operating Conditions
    4. 6.4 Thermal Information
    5. 6.5 Electrical Characteristics
    6. 6.6 Dissipation Ratings
    7. 6.7 Typical Characteristics
  7. Detailed Description
    1. 7.1 Overview
    2. 7.2 Functional Block Diagram
    3. 7.3 Feature Description
      1. 7.3.1 Main Boost Converter
      2. 7.3.2 Power Good Output
      3. 7.3.3 Enable and Power-On Sequencing (EN, ENR)
      4. 7.3.4 Positive Charge Pump
      5. 7.3.5 Negative Charge Pump
      6. 7.3.6 Linear Regulator Controller
      7. 7.3.7 Soft Start
      8. 7.3.8 Fault Protection
      9. 7.3.9 Thermal Shutdown
    4. 7.4 Device Functional Modes
      1. 7.4.1 Enabling and Disabling the Device
  8. Application and Implementation
    1. 8.1 Application Information
    2. 8.2 Typical Application
      1. 8.2.1 Design Requirements
      2. 8.2.2 Detailed Design Procedure
        1. Boost Converter Design Procedure
          1.  Inductor Selection
          2.  Output Capacitor Selection
          3.  Input Capacitor Selection
          4.  Rectifier Diode Selection
          5.  Converter Loop Design and Stability
          6.  Design Procedure Quick Steps
          7.  Setting the Output Voltage and Selecting the Feedforward Capacitor
          8.  Compensation
          9.  Negative Charge Pump
          10. Positive Charge Pump
          11. Linear Regulator Controller
      3. 8.2.3 Application Curves
    3. 8.3 System Examples
  9. Power Supply Recommendations
  10. 10Layout
    1. 10.1 Layout Guidelines
    2. 10.2 Layout Example
    3. 10.3 Thermal Considerations
  11. 11Device and Documentation Support
    1. 11.1 Device Support
      1. 11.1.1 Third-Party Products Disclaimer
    2. 11.2 Documentation Support
      1. 11.2.1 Related Documentation
    3. 11.3 Related Links
    4. 11.4 Receiving Notification of Documentation Updates
    5. 11.5 Community Resources
    6. 11.6 Trademarks
    7. 11.7 Electrostatic Discharge Caution
    8. 11.8 Glossary
  12. 12Mechanical, Packaging, and Orderable Information



  • RGE|24
  • PWP|24

8 Application and Implementation


Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality.

8.1 Application Information

The TPS6514x devices have been designed to provide the input supply voltages for the source drivers and gate drivers in LCD displays. Additionally, they include a linear regulator controller that can be used with an external transistor to provide a regulated 3.3-V output for the digital circuits for LCD panels powered by a 5-V supply rail.

8.2 Typical Application

Figure 3 shows a typical application circuit for a monitor display powered from a 5-V supply. It generates up to 350 mA at 15 V to power the source drivers, and 20 mA at 30 V and –12 V to power the gate drivers.

TPS65140 TPS65141 TPS65145 ai_typicalApp.gif Figure 3. Typical Application Schematic

8.2.1 Design Requirements

Table 1 shows the design parameters for this example.

Table 1. Design Requirements

VI Input supply voltage 2.7 V to 5.8 V
VO1 Boost converter output voltage and current Up to 15 V at 350 mA
VO3 Positive charge pump output voltage and current Up to 30 V at 20 mA
VO2 Negative charge pump output voltage and current Down to –12 V at 20 mA
VO4 Linear regulator controller output voltage and current 3.3 V at 500 mA

8.2.2 Detailed Design Procedure Boost Converter Design Procedure

The first step in the design procedure is to calculate the maximum possible output current of the main boost converter under certain input and output voltage conditions. The following is an example for a 3.3-V to 10-V conversion:

VIN = 3.3 V, VOUT = 10 V, Switch voltage drop VSW = 0.5 V, Schottky diode forward voltage VD = 0.8 V

  1. Duty cycle:
  2. Equation 1. TPS65140 TPS65141 TPS65145 q_dutyc_lvs497.gif
  3. Average inductor current:
  4. Equation 2. TPS65140 TPS65141 TPS65145 q_avg_ind_lvs497.gif
  5. Inductor peak-to-peak ripple current:
  6. Equation 3. TPS65140 TPS65141 TPS65145 q_pp_rip_cur_lvs497.gif
  7. Peak switch current:
  8. Equation 4. TPS65140 TPS65141 TPS65145 q_avg_ind2_lvs497.gif

The integrated switch, the inductor, and the external Schottky diode must be able to handle the peak switch current. The calculated peak switch current must be equal to or lower than the minimum N-MOSFET switch current limit as specified in Electrical Characteristics (1.6 A for the TPS65140 and TPS65141 and 0.96 A for the TPS65145). If the peak switch current is higher, then the converter cannot support the required load current. This calculation must be done for the minimum input voltage where the peak switch current is highest. The calculation includes conduction losses like switch rDS(on) (0.5 V) and diode forward drop voltage losses (0.8 V). Additional switching losses, inductor core and winding losses, and so forth, require a slightly higher peak switch current in the actual application. The above calculation still allows for a good design and component selection. Inductor Selection

Several inductors work with the TPS6514x. Especially with the external compensation, the performance can be adjusted to the specific application requirements. The main parameter for the inductor selection is the saturation current of the inductor which must be higher than the peak switch current as calculated above with additional margin to cover for heavy load transients and extreme start-up conditions. Another method is to choose the inductor with a saturation current at least as high as the minimum switch current limit of 1.6 A for the TPS65140 and TPS65141 and 0.96 A for the TPS65145. The different switch current limits allow selection of a physically smaller inductor when less output current is required. The second important parameter is the inductor DC resistance. Usually, the lower the DC resistance, the higher the efficiency. However, the inductor DC resistance is not the only parameter determining the efficiency. Especially for a boost converter where the inductor is the energy storage element, the type and material of the inductor influences the efficiency as well. Especially at high switching frequencies of 1.6 MHz, inductor core losses, proximity effects, and skin effects become more important. Usually, an inductor with a larger form factor yields higher efficiency. The efficiency difference between different inductors can vary from 2% to 10%. For the TPS6514x, inductor values from 3.3 µH to 6.8 µH are a good choice but other values can be used as well. Possible inductors are shown in Table 2.

Table 2. Inductor Selection

TPS65140 4.7 µH Coilcraft DO1813P-472HC 8.89 × 6.1 × 5 2.6 A, 54 mΩ
4.2 µH Sumida CDRH5D28 4R2 5.7 × 5.7 × 3 2.2 A, 23 mΩ
4.7 µH Sumida CDC5D23 4R7 6 × 6 × 2.5 1.6 A, 48 mΩ
3.3 µH Wuerth Elektronik 744042003 4.8 × 4.8 × 2 1.8 A, 65 mΩ
4.2 µH Sumida CDRH6D12 4R2 6.5 × 6.5 × 1.5 1.8 A, 60 mΩ
3.3 µH Sumida CDRH6D12 3R3 6.5 × 6.5 × 1.5 1.9 A, 50 mΩ
TPS65145 3.3 µH Sumida CDPH4D19 3R3 5.1 × 5.1 × 2 1.5 A, 26 mΩ
3.3 µH Coilcraft DO1606T-332 6.5 × 5.2 × 2 1.4 A, 120 mΩ
3.3 µH Sumida CDRH2D18/HP 3R3 3.2 × 3.2 × 2 1.45 A, 69 mΩ
4.7 µH Wuerth Elektronik 744010004 5.5 × 3.5 × 1 1 A, 260 mΩ
3.3 µH Coilcraft LPO6610-332M 6.6 × 5.5 × 1 1.3 A, 160 mΩ Output Capacitor Selection

For best output voltage filtering, TI recommends a low-ESR output capacitor. Ceramic capacitors have a low ESR value but depending on the application, tantalum capacitors can be used as well. A 22-µF ceramic output capacitor works for most of the applications. Higher capacitor values can be used to improve load transient regulation. See Table 2 for the selection of the output capacitor. The output voltage ripple can be calculated as:

Equation 5. TPS65140 TPS65141 TPS65145 q_ou_cap_sel_lvs497.gif


  • IP = Peak switch current as calculated in the previous section with ISW(peak)
  • L = Selected inductor value
  • IOUT = Normal load current
  • fS = Switching frequency
  • VD = Rectifier diode forward voltage (typical 0.3 V)
  • COUT = Selected output capacitor
  • ESR = Output capacitor ESR value Input Capacitor Selection

For good input voltage filtering, TI recommends low-ESR ceramic capacitors. A 22-µF ceramic input capacitor is sufficient for most of applications. For better input voltage filtering, this value can be increased. See Table 3 and the typical applications for input capacitor recommendations.

Table 3. Input and Output Capacitors Selection

22 µF, 1210 16 V Taiyo Yuden EMK325BY226MM CO
22 µF, 1206 6.3 V Taiyo Yuden JMK316BJ226 CI Rectifier Diode Selection

To achieve high efficiency, a Schottky diode must be used. The voltage rating must be higher than the maximum output voltage of the converter. The average forward current must be equal to the average inductor current of the converter. The main parameter influencing the efficiency of the converter is the forward voltage and the reverse leakage current of the diode; both must be as low as possible. Possible diodes are: On Semiconductor MBRM120L, Microsemi UPS120E, and Fairchild Semiconductor MBRS130L. Converter Loop Design and Stability

The TPS6514x converter loop can be externally compensated and allows access to the internal transconductance error amplifier output at the COMP pin. A small feedforward capacitor across the upper feedback resistor divider speeds up the circuit as well. To test the converter stability and load transient performance of the converter, a load step from 50 mA to 250 mA is applied, and the output voltage of the converter is monitored. Applying load steps to the converter output is a good tool to judge the stability of such a boost converter. Design Procedure Quick Steps

  1. Select the feedback resistor divider to set the output voltage.
  2. Select the feedforward capacitor to place a zero at 50 kHz.
  3. Select the compensation capacitor on pin COMP. The smaller the value, the higher the low frequency gain.
  4. Use a 50-kΩ potentiometer in series to CC and monitor VOUT during load transients. Fine tune the load transient by adjusting the potentiometer. Select a resistor value that comes closest to the potentiometer resistor value. This must be done at the highest Vin and highest load current because stability is most critical at these conditions. Setting the Output Voltage and Selecting the Feedforward Capacitor

The output voltage is set by the external resistor divider and is calculated as:

Equation 6. TPS65140 TPS65141 TPS65145 q_set_out_v_lvs497.gif

Across the upper resistor, a bypass capacitor is required to speed up the circuit during load transients as shown in Figure 4.

TPS65140 TPS65141 TPS65145 Schem_02_SLVS497.gif Figure 4. Feedforward Capacitor

Together with R1 the bypass capacitor C8 sets a zero in the control loop at approximately 50 kHz:

Equation 7. TPS65140 TPS65141 TPS65145 q_bypass_lvs497.gif

A value closest to the calculated value must be used. Larger feedforward capacitor values reduce the load regulation of the converter and cause load steps as shown in Figure 5.

TPS65140 TPS65141 TPS65145 scp_ld_st_lvs497.gif Figure 5. Load Step Caused by a Too Large Feedforward Capacitor Value Compensation

The regulator loop can be compensated by adjusting the external components connected to the COMP pin. The COMP pin is connected to the output of the internal transconductance error amplifier. A typical compensation scheme is shown in Figure 6.

TPS65140 TPS65141 TPS65145 Schem_03_SLVS497.gif Figure 6. Compensation Network

The compensation capacitor CC adjusts the low frequency gain, and the resistor value adjusts the high frequency gain. The following formula calculates at what frequency the resistor increases the high frequency gain.

Equation 8. TPS65140 TPS65141 TPS65145 q_rccp_lvs497.gif

Lower input voltages require a higher gain and a lower compensation capacitor value. A good start is CC = 1 nF for a 3.3-V input and CC = 2.2 nF for a 5-V input. If the device operates over the entire input voltage range from 2.7 V to 5.8 V, TI recommends a larger compensation capacitor up to 10 nF. Figure 7 shows the load transient with a larger compensation capacitor, and Figure 8 shows a smaller compensation capacitor.

TPS65140 TPS65141 TPS65145 scp_lg_cap_lvs497.gif Figure 7. CC = 4.7 nF
TPS65140 TPS65141 TPS65145 scp_sm_cap_lvs497.gif Figure 8. CC = 1 nF

Lastly, RC must be selected. A good practice is to use a 50-kΩ potentiometer and adjust the potentiometer for the best load transient where no oscillations should occur. These tests have to be done at the highest VIN and highest load current because the converter stability is most critical under these conditions. Figure 9, Figure 10, and Figure 11 show the fine tuning of the loop with RC.

TPS65140 TPS65141 TPS65145 scp_ovrcomp_lvs497.gif Figure 9. Overcompensated (Damped Oscillation),
RC is Too Large
TPS65140 TPS65141 TPS65145 scp_optm_rc_lvs497.gif Figure 11. Optimum, RC is Ideal
TPS65140 TPS65141 TPS65145 scp_udrcomp_lvs497.gif Figure 10. Undercompensated (Loop is Too Slow),
RC is Too Small Negative Charge Pump

The negative charge pump provides a regulated output voltage by inverting the main output voltage, VO1. The negative charge pump output voltage is set with external feedback resistors.

The maximum load current of the negative charge pump depends on the voltage drop across the external Schottky diodes, the internal ON-resistance of the charge pump MOSFETS Q8 and Q9, and the impedance of the flying capacitor, C12. When the voltage drop across these components is larger than the voltage difference from VO1 to VO2, the charge pump is in drop out, providing the maximum possible output current. Therefore, the higher the voltage difference between VO1 and VO2, the higher the possible load current. See Figure 21 for the possible output current versus boost converter voltage VO1 and the calculations below.

Equation 9. VOUTmin = –(VO1 – 2 VF – IO (2 × rDS(on)Q8 + 2 × rDS(on)Q9 + Xcfly))

Setting the output voltage:

Equation 10. TPS65140 TPS65141 TPS65145 q_vo_ch2_lvs497.gif
Equation 11. TPS65140 TPS65141 TPS65145 q_r3_neg_lvs497.gif

The lower feedback resistor value, R4, must be in a range from 40 kΩ to 120 kΩ or the overall feedback resistance must be within 500 kΩ to 1 MΩ. Smaller values load the reference too heavy and larger values may cause stability problems. The negative charge pump requires two external Schottky diodes. The peak current rating of the Schottky diode must be twice the load current of the output. For a 20-mA output current, the dual Schottky diode BAT54 or similar is a good choice. Positive Charge Pump

The positive charge pump can be operated in a voltage doubler mode or a voltage tripler mode depending on the configuration of the C2+ and C2–/MODE pins. Leaving the C2+ pin open and connecting C2-/MODE to GND forces the positive charge pump to operate in a voltage doubler mode. If higher output voltages are required the positive charge pump can be operated as a voltage tripler. To operate the charge pump in the voltage tripler mode, a flying capacitor must be connected to C2+ and C2–/MODE.

The maximum load current of the positive charge pump depends on the voltage drop across the internal Schottky diodes, the internal ON-resistance of the charge pump MOSFETS, and the impedance of the flying capacitor. When the voltage drop across these components is larger than the voltage difference VO1 × 2 to VO3 (doubler mode) or VO1 × 3 to VO3 (tripler mode), then the charge pump is in dropout, providing the maximum possible output current. Therefore, the higher the voltage difference between VO1 x 2 (doubler) or VO1 × 3 (tripler) to VO3, the higher the possible load current. See Figure 22 and Figure 23 for output current versus boost converter voltage, VO1, and the following calculations.

Voltage doubler:

Equation 12. VO3max = 2 × VO1 – (2 VF + 2 × IO × (2 × rDS(on)Q5 + rDS(on)Q3 + rDS(on)Q4 + XC1))

Voltage tripler:

Equation 13. VO3max = 3 × VO – (4 × VF + 2 × IO × (3 × rDS(on)Q5 + rDS(on)Q3 + rDS(on)Q4 + XC1 + XC2))

The output voltage is set by the external resistor divider and is calculated as:

Equation 14. TPS65140 TPS65141 TPS65145 q_ext_res_lvs497.gif
Equation 15. TPS65140 TPS65141 TPS65145 q_ext_res2_lvs497.gif Linear Regulator Controller

The TPS6514x includes a linear regulator controller to generate a 3.3-V rail when the system is powered from a 5-V supply. Because an external NPN transistor is required, the input voltage of the TPS6514x applied to VIN must be higher than the output voltage of the regulator. To provide a minimum base drive current of 13.5 mA, a minimum internal voltage drop of 500 mV from VI to VBASE is required. This can be translated into a minimum input voltage on VIN for a certain output voltage as the following calculation shows:

Equation 16. VI(min) = VO4 + VBE + 0.5 V

The base drive current together with the hFE of the external transistor determines the possible output current. Using a standard NPN transistor like the BCP68 allows an output current of 1 A and using the BCP54 allows a load current of 337 mA for an input voltage of 5 V. Other transistors can be used as well, depending on the required output current, power dissipation, and PCB space. The device is stable with a 4.7-µF ceramic output capacitor. Larger output capacitor values can be used to improve the load transient response when higher load currents are required.

8.2.3 Application Curves

TPS65140 TPS65141 TPS65145 eff_v_il_lvs497.gif Figure 12. Efficiency vs Load Current
TPS65140 TPS65141 TPS65145 eff_v_vi_lvs497.gif Figure 14. Efficiency vs Input Voltage
TPS65140 TPS65141 TPS65145 pwm_dscon_mo_lvs497.gif Figure 16. PWM Operation at Light Load
TPS65140 TPS65141 TPS65145 ld2_trn_res_lvs497.gif Figure 18. Load Transient Response
TPS65140 TPS65141 TPS65145 sfst_vo1_lvs497.gif Figure 20. Soft Start VO1
TPS65140 TPS65141 TPS65145 vo3_max_ld_lvs497.gif Figure 22. VO3 Maximum Load Current
TPS65140 TPS65141 TPS65145 eff2_v_il_lvs497.gif Figure 13. Efficiency vs Load Current
TPS65140 TPS65141 TPS65145 pwm_con_mo_lvs497.gif Figure 15. PWM Operation Continuous Mode
TPS65140 TPS65141 TPS65145 ld_trn_res_lvs497.gif Figure 17. Load Transient Response
TPS65140 TPS65141 TPS65145 pwr_up_lvs497.gif Figure 19. Power-Up Sequencing
TPS65140 TPS65141 TPS65145 vo2_max_l_lvs497.gif Figure 21. VO2 Maximum Load Current
TPS65140 TPS65141 TPS65145 vo3-2_max_ld_lvs497.gif Figure 23. VO3 Maximum Load Current

8.3 System Examples

TPS65140 TPS65141 TPS65145 ai_notebook_SLVS497.gif Figure 24. Typical Application, Notebook Supply
TPS65140 TPS65141 TPS65145 ai_monitor_SLVS497.gif Figure 25. Typical Application, Monitor Supply